Receiver architectures for frequency-modulated signals or FM signals, respectively, based on a limiting circuit and using digital frequency modulators are known, where an intermediate frequency (IF) is used, which is close to the sampling frequency of the receiver architecture. Such a procedure, where the sampling frequency differs only slightly from a frequency of a signal to be sampled, is referred to as subsampling and can only be applied in a few selected cases of frequency constellations. For the present case of a receiver architecture for frequency-modulated signals based on a limiting circuit and digital frequency-demodulator circuit, which operates at a sampling frequency close to the intermediate frequency, this means that the circuit can only be used in very limited frequency constellations in the intermediate frequency and the sampling frequency.
FIG. 4 shows a digital FM demodulation circuit 800, which combines the concept of subsampling with a delay and multiply FM demodulator. Such a solution is, for example, implemented in the electronic device TDA5230. Here, the circuit 800 has an analog input circuit 810 in an HF range with an input 810a for an HF signal, and an output 810b for an IF signal, wherein HF stands for high frequency and IF for intermediate frequency. The IF signal provided at the output 810b of the analog input circuit 810 is supplied to an input of an IF filter 820 in the IF range of the circuit 800. An output of the IF filter 820 is connected to an input of a limiting circuit 830. Thus, a connection, which connects an output of the limiting circuit 830 to an input of a sampling means 850, which is also referred to as sampler in FIG. 4, carries a one bit data stream with a frequency corresponding to the intermediate frequency of the IF signal. Apart from the sampling means 850, a sampling and digital demodulation range of the signal processing circuit 800 comprises a so-called zero IF mixer 860 and a filter and demodulation circuit 870. Thereby, the zero IF mixer 860 is connected between the output of the sampling means 850 and the first and second input of the filter and demodulation circuit 870, which provides a non-phase-shifted output signal (I path) and an output signal shifted by 90° (Q path) of the zero IF mixer 860 of the filter and demodulation circuit 870. The zero IF mixer 860 has a first internal IQ mixer 880-1, and a second internal IQ mixer 880-2, which are connected in series, a sine-wave generator 890 and a numerical controlled oscillator 900. The filter and demodulation circuit 870 has a first CIC filter 910-1 for the I path and a second CIC filter 910-2 for the Q path as low-pass filter, as well as a delay and multiply demodulator 920. Thus, a sampling and digital demodulation range comprises the sampler 850, the zero IF mixer 860 and the filter and demodulation circuit 870.
Below, the mode of operation of the signal processing circuit 800 will be briefly discussed. An HF signal supplied to the input 810a of the analog input circuit 810 is downconverted by the input circuit 810 to the IF signal with an intermediate frequency, which is smaller than the frequency of the HF signal. The IF signal provided at the output 810b of the input circuit 810 is transformed into a signal derived from the IF signal in the IF range of the signal processing circuit 800, which means by the IF filter 820 and the limiting circuit 830. The output signal of the limiter 830 is referred to as 1 bit data stream@IF in FIG. 4. This output signal is then sampled with a sampling frequency fs in the sampling means 850, and supplied to the zero IF mixer and the filter and demodulation circuit 870 for further processing. The sampling frequency fs is thereby by a factor k higher than the intermediate frequency IF. Thus, the sampling frequency fs and the intermediate frequency fulfill the relationfs=k·IF. 
As has already been mentioned, the signal processing circuit 800 is based on the principle of the subsampling approach. By subsampling, the prior art and the present invention mean that the sampling frequency fs is smaller than the double of a signal frequency, or in the present case the frequency of the IF signal IF, respectively, which means that the numerical factor k is <2, so that the sampling theorem or the Nyquist-Shannon sampling theorem is violated.
The two IQ mixers 880-1 and 880-2 of the zero IF mixer 860 serve to mix the output signal of the sampling means 850 to approximately zero with regard to the frequency. For that purpose, a first mixing frequency is provided to the first IQ mixer 880-1 by the sine-wave generator, which corresponds to a quarter of the sampling frequency or system frequency fs, respectively. The final mixing to approximately zero is obtained with the second IQ mixer 880-2, to which a second mixing frequency is provided by the numerical controlled oscillator 900. The two output signals of the zero IF mixer 860 are filtered by the two CIC filters 910-1 and 910-2 in the filter and demodulation circuit 870. Then, demodulation is performed in the delay and multiply demodulator 920.
Due to the fact that a square wave signal has all odd-numbered harmonics or harmonic waves apart from the fundamental wave, the sampling function, which is effected by the sampling means 850, convolves the fundamental wave with all odd-numbered harmonics of the output signal of the limiting circuit 830.
This method, which means the method of subsampling, functions only in special circumstances, when a special relation between the sampling frequency of the sampling means 850 and the signal frequency, which means in this case the frequency of the digital IF signal fIF, is fulfilled, because only in this case a sufficient distance can be maintained in the frequency range between the fundamental wave and the higher harmonics after convolution, which is a consequence of the sampling process.
Special relations between the sampling frequency and the intermediate frequency occur across a large range of sampling frequencies and intermediate frequencies. In this case, it can happen that the higher harmonics of the digital IF signal are very close to the fundamental wave, or, in the worst case, even have the same frequency, respectively, after sampling in the frequency range. This can result in a heavy distortion of the following demodulation of the signal and thus a high bit error rate, for example in the case of FSK signal detection (frequency shift keying). Thus, the reception of a data stream can even be made completely impossible.
FIG. 5 shows a frequency distribution or frequency plan, respectively, of signals occurring in the circuit of FIG. 4. FIG. 5a shows the frequency distribution of the IF signal at the output of the limiting circuit or the output of the limiter 830, respectively. In the following, the signal at the output of limiter 830 in FIG. 5 is also referred to as LIM. FIG. 5b shows the frequency distribution of the sampling function. FIG. 5c shows the frequency distribution, which results by the convolution of the signal LIM with the sampling function (sampling FKT) resulting during sampling. FIG. 5d shows the frequency distribution of a signal, which the sine-wave generator 890 provides to the first internal mixer 880-1 of the zero IF mixer 860. This signal has a frequency, which corresponds to a quarter of the sampling frequency fs. FIG. 5e shows the frequency distribution of a signal at an output of the first internal mixer 880-1 of the zero IF mixer 860, wherein the frequency distribution is a result of mathematical convolution of the signal at the output of the sampling means 850, and a multiplication of the signal of the sine-wave generator 890 with the frequency fs/4.
In other words, FIG. 5 shows the frequency distribution or the frequency plan, respectively, of the fundamental wave and the third harmonic, which represents the first odd-numbered harmonic of the digital IF signal, after sampling and mixing in the first stage of the zero IF mixer 860. The five frequency axes of the samples 5a to 5e are scaled identically and are illustrated without being displaced to each other. Both the positions and the distances on the frequency axis can thus be transferred by a vertical displacement between the partial emitters. In other words, the distances between the third harmonics or the first odd-numbered harmonic and the fundamental wave are defined in relation to the sampling frequency and the intermediate frequency of the IF signal.
Based on the convolution of the sampling function, which is referred to as sample FKT in FIG. 5, and the IF signal, the frequency distribution of the signal at the output of the sampling means 850 shown in FIG. 5c results. In the case of subsampling as it is used here, the frequency distribution has, for example, a contribution at a frequency (fs−Zf) at the output of the sampling means 850, which originates from the contribution of the digital IF signal at the frequency Zf and the contribution of the sampling function at the frequency fs, and a contribution at the frequency (3Zf−2fs) which originates from the first odd-numbered harmonic of the digital IF signal with frequency 3Zf and the contribution of the sampling function at a frequency of 2fs. These two contributions in the frequency distribution of the signal at the output of the sampling means 850 are indicated with the reference number 950 in FIG. 5c, since they represent a critical range in the frequency band for the demodulation of the signal within an FSK signal detection (frequency shift keying) following the demodulation, since the intermediate frequency and the sampling frequency meet in a critical range. As has already been explained above, due to the low distance of the two contributions, heavy distortions can result in the demodulation following the sampling which can finally cause a high bit error rate.
Earlier solutions resulted in a very high hardware and development effort, which is reflected in high chip area requirements and high power consumption.